The Sorensen Audio Experiment, design description version 2.1
Contents
3.3. Error
Amplifier / Integrator
3.5. Voltage
swing increase circuit
4. Implementation Considerations
5. Known limitations / issues for future work
1.Copyright Notice
Copyright (c) 2001 Johan Sörensen.
Permission is granted to copy,
distribute and/or modify this document
under the terms of the GNU Free
Documentation License, Version 1.1
or any later version published by the
Free Software Foundation;
with the Invariant Sections being
section 1. (this copyright notice), with no
Front-Cover Texts, and with no
Back-Cover Texts.
A copy of the license is included in the
file entitled "License.htm".
2.Introduction
The
original aim with this design was to realize a High Fidelity audio power
amplifier, in a simple, and relatively cheap way. That is, cheap compared to
other commercial High End audio equipment. The result is probably not comparable
to true High End amplifiers, but it’s certainly closer to High End, than to the
Low End integrated systems you buy in consumer electronics stores at discount
prices.
The design
operates in class D, since the output transistors alternate between only two
states; completely off and completely on. They alternate between these two
states many times per second, creating a pulse-width modulated output signal.
This is then filtered, to closely reproduce the desired music signal in the
audible range.
There are
different ways of creating a pulse-width modulated signal. A common one is to
compare an analog input signal with a high-frequency triangle wave, and drive
the output to high when the input is higher than the momentary triangle wave
voltage, and drive the output to low otherwise. This creates a pulse-width
modulated output signal, with a switch frequency equal to the frequency of the
triangle wave.
This design
is slightly different. The picture below shows the generalized elements of the
circuit.
This
circuit can be described as a comparator driving an output stage (to either
“high” or “low” state), with a feedback loop around the whole thing, and a
noise-shaping loop filter providing the input to the comparator. A theoretical
analysis of the noise-shaping filter can be found at http://listen.to/audioexperiment.
There is
nothing setting a fixed switch frequency in this picture. If all components
were ideal, the switch frequency would be virtually infinite. But in reality,
components are far from ideal (comparators have lag-time, transistors take time
to switch on and off, and so on…). This means that all the components in the
feedback loop in this design together sets the switch frequency. This is
further explained below.
3.Circuit Description
3.1.Overview
The
descriptions in this chapter refer to the schematics
diagram.
The present
design described in this document creates the pulse-width modulated output
signal through self-oscillation. Oscillation is something that is normally
avoided at all cost in amplifiers in general, but this case is very different.
In this
design, the comparator, the voltage swing increase circuit, and the power
output stage, together make a very high gain amplifier, which is (only)
optimized to swing to the positive or negative supply voltage alternatively. A
feedback loop is then applied around this composite amplifier, and an
integrating error amplifier controls the input to the aforementioned composite
amplifier. This in effect creates a very fast on-off regulation, which in real
life oscillates at 1.5 ~ 2 MHz. This pulse-width modulation frequency is
significantly higher than the highest audible frequency to any human.
Another way
of understanding the operation of this circuit is that the error amplifier
compares the integrated momentary value of the output stage with the desired value
defined by the input signal, and then drives the output to the positive or
negative supply respectively depending on the outcome of the comparison. Of
course, driving the output to the positive supply for a while invariably and
quickly leads to a integrated output value that is slightly “too high”, thus
forcing the output to the negative supply instead, and so on… The point is that
this process is so quick and accurate compared to the fluctuations in the input
signal, that the latter is followed quite closely by the filtered output
signal.
The
remaining sections in this chapter contain some notes and explanations of the
different sub-circuits of the complete design.
3.2.Input stage
The input
stage is just there to provide an impedance adjustment to the relatively low
input impedance of the next stage (error amplifier/integrator). In the
schematics, this is an LT1056 operational amplifier (U12) – however there are
of course alternatives to this. Depending on preferences, and price sensitivity,
this part can be replaced by any operational amplifier that is stable at unity
gain.
Or, one
bipolar junction transistor, in a voltage follower configuration, could do.
However, be aware that a transistor voltage follower is not perfectly linear,
and thus introduces some unnecessary (and audible) distortion. (So does cheap
operational amplifiers J.)
3.3.Error Amplifier / Integrator
The LF356
(U10) serves as an integrating error amplifier. (The integration is a critical
part, in that it does noise shaping.)
The value
of C8 determines the integration constant. The suggested value of C8 has been
determined empirically – you may try to tweak this up or down slightly, and
observe the effect on the overall amplifier operation (in particular, the audible
result in terms of distortion).
3.4.Comparator
The
comparator used is the LM311 (U11). This component unfortunately only has an
open collector output, which means that it’s output can only pull to the negative
supply, and there has to be a pull-up resistor to the positive supply. To
alleviate the bad positive drive capability, a BF245A JFET transistor (J2) is
applied directly after the U11 output. Since J2 is configured as a voltage
follower, it in turn has a not-so-good negative pull capability (due to R25).
This is finally alleviated by D12-R26, such that the final output of this stage
(the Source on J2), has a good positive as well as negative drive capability.
3.5.Voltage swing increase circuit
Q1-Q4 make
up the voltage swing increase circuit, increasing voltage switch of the PWM
output from ±5 V to whatever supply voltage is supplied over Q3-Q4. (With the
components used, there is a limit of some ±30V for what voltage this circuit
can withstand). This circuit is quite fast; a delay of only some ten
nanoseconds is measurable between input and output of this circuit.
C1-C2 are
feed-forward capacitors, turning on and off Q3 and Q4 alternatively, such that
only one of Q3 or Q4 should conduct at any time. However, there seems to be a
slight overlap, and as the supply voltages to this part are increased, Q3-Q4
tend to warm up, if D1 isn’t there to introduce a small delay/resistance
between Q3 and Q4.
3.6.Power output stage
The output
transistors NTE54/NTE55 (Q7-Q8) don’t have enough current amplification
capability (Hfe) to be directly driven from the output of the
voltage swing increase circuit (Q1-Q4). This is why there is an extra stage in
between; Q5-Q6.
3.7.Output filter
This is a
part that I haven’t been able to design and test properly, due to a PCB layout
mistake in my own builds of this design (also see the next chapter). So I
admittedly have a problem with radio frequency noise being emitted from the
speaker wires, creating a problem to receive FM radio with an antenna placed
too close to the power amplifier L.
4.Implementation Considerations
There are
issues relating to the PCB design implementation and output filtering, which
are important for the overall performance in a real embodiment of the design.
Be careful how you draw the power supply and earth strips, considering where
high frequency currents will flow. The abstract schematics design does not
reflect the capacitor bypassing needed in various places for successful
operation J.
Create one
ground for the Low Voltage parts (the ones supplied with ± 5V), and one ground for the High Voltage parts. Then
pull wires from these two grounds to a common ground point at the power supply.
In the
output filter, the ground side of C5 must be connected to the High Voltage
ground; this is the same ground point you use for bypassing the ± 25V supplies with capacitors of at least a
few hundred µF. Also, bypass the supplies with 100nF capacitors, with really
good high frequency characteristics. (As close to the output transistors as
possible.)
5.Known limitations / issues for
future work
- The major limiting factor for higher
output powers is the voltage tolerance of the power output stage and the
voltage swing increase circuit. In particular, the latter circuit (Q1-Q4),
tends to get too warm and break down if the supply voltage exceeds ± 30
Volt.
- The fact that the bipolar output power
transistors are connected in a common-emitter follower configuration means
that some voltage is lost on the actual output, compared to the output of
the voltage swing increase circuit (collector of Q4). This of course leads
to some undesirable power dissipation in Q5-Q8.
Feeding the voltage swing increase circuit with a couple of volts higher supply voltage can possibly alleviate this issue, but this also leads to a more complex power supply… - The LM311 comparator is not ideal, since
it has an open collector output, and thus has no positive drive capability
in itself. (Which is countered for
with J2-D12.)
- Of course, it would be better and cheaper
not to need a impedance buffer on the input (U12). The values of R22/R27
determine the overall amplification factor, but they also have a
significant impact on amplifier performance, and ultimately, stability.
Perhaps R27 can be increased, but I couldn’t get the circuit to work well
with a R27 value in the range that I would consider necessary (~20kohm) in
order to not need the impedance buffer.
- Output filtering to reduce radio frequency
noise emission will be a critical issue for any future class D design
allowing higher output power to the speakers.